Reference generation circuit for maintaining temperature-tracked linearity in amplifier with adjustable high-frequency gain

ABSTRACT

Equalizing an input signal according to a receiver equalizer peaking circuit having a capacitor FET (CFET) providing a capacitive value and a resistor FET (RFET) providing a resistive value, generating a capacitor control voltage at a gate of the CFET using a capacitor controller DAC based on a first reference voltage, and a RFET control voltage at a gate of the RFET using a resistor controller DAC based on a second reference voltage, generating the first reference voltage using a replica input FET, the first reference voltage varying according to a threshold voltage (Vt) of an input FET, providing the first reference voltage to the capacitor controller DAC, generating the second reference voltage using a replica RFET, the second reference voltage varying with respect to the first reference voltage and a Vt of the replica of the RFET, and providing the second reference voltage to the resistor controller DAC.

REFERENCES

The following prior applications are herein incorporated by reference intheir entirety for all purposes:

U.S. application Ser. No. 16/378,461, filed Apr. 8, 2019, naming SuhasRattan, entitled “Amplifier with Adjustable High-Frequency Gain UsingVaractor Diodes”, hereinafter referred to as [Rattan I].

BRIEF DESCRIPTION

Methods and systems are described for equalizing an input signalaccording to a receiver equalizer peaking circuit having a capacitor FET(CFET) providing a capacitive value and a resistor FET (RFET) providinga resistive value, generating a capacitor control voltage at a gate ofthe CFET using a capacitor controller DAC based on a first referencevoltage, and a RFET control voltage at a gate of the RFET using aresistor controller DAC based on a second reference voltage, generatingthe first reference voltage using a replica input FET, the firstreference voltage varying according to a threshold voltage (Vt) of aninput FET, providing the first reference voltage to the capacitorcontroller DAC to maintain temperature-tracked DAC linearity, generatingthe second reference voltage using a replica RFET, the second referencevoltage varying with respect to (i) the first reference voltage and (ii)a Vt of the replica of the RFET, and providing the second referencevoltage to the resistor controller DAC to maintain thetemperature-tracked DAC linearity.

Furthermore, a controller circuit providing bias to an amplifier circuitincorporating configurable frequency compensation is described, as wellas a reference generation circuit providing reference voltages to thecontroller circuit, the combination being suitable for use as acontinuous-time linear equalizer (CTLE) for communications receiverinput signals. Elements of the design minimize behavioral variation overprocess, voltage, and temperature variation, while facilitating compactcircuit layout with the configurable elements closely integrated withthe analog devices they control.

BRIEF DESCRIPTION OF FIGURES

FIG. 1 is a circuit diagram of a CTLE amplifier.

FIG. 2 is a simplified circuit diagram of a CTLE amplifier as in FIG. 2,showing two DAC adjustment controls.

FIG. 3 is a circuit diagram providing pre-compensated bias voltages to aCTLE circuit.

FIG. 4 is a flowchart of a method 400, in accordance with someembodiments.

DETAILED DESCRIPTION

Continuous-time Linear Equalization (CTLE) circuits are well known inthe art. One common design is based on a conventional differentialamplifier circuit utilizing a matched transistor pair having individualsource loads but common drain connections to a fixed current sink.Splitting the current sink into two, one for each transistor drain,allows the drains to be cross-coupled with a frequency-dependentimpedance such as a parallel RC network, modifying the essentially flatgain-vs-frequency characteristic of the basic differential amplifierinto one having distinctly different low- and high-frequency gains.

In communications system receivers, such a CTLE circuit is typicallyconfigured to provide increased high-frequency gain to equalize orcompensate for the inevitable high frequency loss of most communicationsmedia. In some embodiments, careful configuration of amplitude andequalization functions is performed to facilitate accurate signaldetection and/or clock recovery by subsequent circuits. In someembodiments, a CTLE circuit in which both the gain characteristics andthe frequency break points of such frequency-dependent compensation maybe adjusted or configured.

Such CTLE circuits are intended for use in an integrated circuitenvironment processing extremely high frequency signals with minimalpower consumption. The available power rails Vdd and Vss may typicallyprovide one volt or less of operating voltage, thus microampere currentflows imply path impedances of many thousands to millions of ohms. Asresistances of these magnitudes may occupy substantial surface area insome integrated circuit processes, active circuit elements such as fieldeffect transistors (FETs) may be preferable to passive elementembodiments. Thus, as representative examples, the CTLE circuit of FIG.1 as described by [Rattan I] incorporates MOS transistor 131 in whichthe channel resistance is used primarily as a source degenerationresistor, or resistor FET (RFET). Similarly, capacitive elements 133 and134 are so-called varactor devices, although such elements are nowtypically implemented using the voltage-dependent gate-to-channelcapacitance of a MOS transistor acting as a source degenerationcapacitor (or capacitor FET, CFET) as an alternative to the historicallyoriginal PN junction. Adjustment of gate voltage permits the effectivechannel resistance of such an RFET or body capacitance of such a CFET tobe modified, facilitating control of the circuit's frequency responsecharacteristics.

The configurable CTLE circuit of FIG. 1 incorporates three degrees ofadjustment capability, as overall gain may be adjusted by configuringthe effective value of load resistance RL, the amount of high-frequencypeaking may be adjusted by configuring the resistance of RFET 131, andthe transition frequency at which high-frequency peaking begins may beadjusted by configuring the capacitance of CFETs 133 and 134. FIG. 1shows that RL may be adjusted in discrete steps based on the number ofparallel resistor elements enabled in the circuit, with the relativechange occurring in each such adjustment step being primarily defined bycircuit design, being only incidentally impacted by the effects ofcircuit voltage, temperature, and/or integrated circuit processvariation.

In contrast, configuration of peaking amplitude and peaking transitionfrequency are implemented using inherently non-linear elements, gatevoltage controlling MOS transistor channel resistance in the first case,and gate voltage controlling MOS transistor base-to-channel capacitancein the second case. It is well understood that the voltage-dependentbody capacitance of MOS transistor devices is both non-linear and can bea function of time and general system characteristics, depending on themanufacturing process used and variations in operating temperatures. Forexample, charge density in active devices changes over time, with thiseffect being much more noticeable in small channel length devices.Similarly, the channel resistance of a MOS device also varies with notonly gate voltage, but also temperature and process variations thatmodify the threshold voltage and other characteristics. Such operationalor parametric variation with changes to supply voltage, operatingtemperature, and/or across device instances presenting variations in theintegrated circuit process are herein collectively described as PVT(e.g. Process, Voltage, and Temperature) variation.

One may observe that even though the control voltages used to configurepeaking amplitude and frequency are provided by, as one example,digital-to-analog converters delivering equal-sized voltage steps, theactual parametric change obtained by each such step will in practice beboth non-linear and PVT dependent. Embodiments herein describe methodsand systems for generating a first reference voltage vreg_cdeg thatmaintains temperature-tracked DAC linearity such that any capacitorcontroller digital-to-analog-converter (DAC) code provides a CFETcontrol voltage that varies responsive to PVT yet maintains a desiredcapacitive value. Similarly, the methods and systems generate a secondreference voltage vreg_rdeg that maintains temperature-tracked DAClinearity such that any resistor controller DAC code provides a RFETcontrol voltage that varies responsive to PVT yet maintains a desiredresistive value.

Compensation for PVT variations generally eschews absolute settings(e.g. a fixed bias to set the transistor operating point) forratiometric ones that take advantage of the close matching of identicaltransistors with each other on the same integrated circuit die, eventhough their absolute functional parameters may vary widely overvoltage, temperature, and process variations. Thus the basicamplification characteristics of the CTLE circuit of FIG. 1 areconsistent and stable, due to the close matching of elements 111, 112,113, to 121, 122, 123, even if, as one example, the exact currentthrough 113/123 is not known. As shown herein, such ratiometricinsensitivity to PVT may be incorporated into the other parametriccontrols of a CTLE.

As shown in FIG. 4, a flowchart of a method 400 is described herein,which may include equalizing 402 an input signal according to a receiverequalizer peaking circuit having a capacitor FET (CFET) providing acapacitive value and a resistor FET (RFET) providing a resistive value.A capacitor control voltage is generated 404 at a gate of the CFET221/222 using a capacitor controller DAC 220 based on a first referencevoltage vreg_cdeg, and a RFET control voltage at a gate of the RFET231/232 using a resistor controller DAC 230 based on a second referencevoltage vreg_rdeg. The first reference voltage is generated 406 using areplica input FET 311, where the first reference voltage varyingaccording to a threshold voltage (Vt) of an input FET 241/242. The firstreference voltage vreg_cdeg is provided 408 to the capacitor controllerDAC 220 to maintain temperature-tracked DAC linearity such that a givencapacitor controller DAC code provides a corresponding CFET controlvoltage that maintains a desired capacitive value of the CFET responsiveto variations in e.g., temperature and/or common mode input. The secondreference voltage vreg_rdeg is generated 410 using a replica RFET 351,the second reference voltage varying with respect to (i) the firstreference voltage vreg_cdeg and (ii) variation in the threshold voltageVt of the replica RFET 351. The second reference voltage is provided 412to the resistor controller DAC 230 to maintain the temperature-trackedDAC linearity.

The CTLE circuit of FIG. 2 explicitly shows adjustment of the RFETcircuit composed of FETs 231/232 and CFET circuit composed of FETs221/222 elements as being controlled by resistor controller andcapacitor controller DACs 230 and 220 respectively. No limitation isimplied, with subsequent references to control, compensation, oradjustment of elements of FIG. 2 being equally applicable to other CTLEcircuits and/or controlling elements, in particular both the NMOS andPMOS designs of [Rattan I].

FIG. 2 is a schematic of a CTLE having a receiver equalizer peakingcircuit that includes a CFET (shown as varactor connected diodes 221 and222) providing a capacitive value and an RFET (shown as a sourcedegeneration switch composed of active FETs 231 and 232) providing aresistive value. As shown, the capacitive and resistive values arecross-coupled between tail nodes 225 and 226 of a differential amplifiercircuit and are adjustable to configure high-frequency peakingcharacteristics of the equalizer peaking circuit. Furthermore, FIG. 2includes a controller circuit composed of a capacitor controller DAC 220configured to provide a CFET control voltage to the DFET based on afirst reference voltage vreg_cdeg and a resistor controller DAC 230configured to provide an RFET control voltage to a gate of the RFETbased on a second reference voltage vreg_rdeg. The first and secondreference voltages may be generated using a reference generationcircuit, described in further detail below with respect to FIG. 3. Nolimitation is implied, with subsequent references to control,compensation, or adjustment of elements of FIG. 2 being equallyapplicable to other CTLE circuits and/or controlling elements, inparticular both the NMOS and PMOS designs of [Rattan I].

As shown in FIG. 2, the gate-to-channel voltage that adjusts theeffective body capacitance of varactors 221 and 222 is a function of thetail node voltage on tail nodes 225 and 226. As transistors used invaractor mode are generally kept in their channel-closed operatingregion, the turn-on threshold voltage Vt of transistors 221 and 222 isnot relevant, thus the PVT compensation of the reference voltageprovided to capacitor controller DAC 220 tracks tail node 225/226voltage variations associated with variations in the Vt of inputtransistors 241 and 242. The reference generation circuit 300 of FIG. 3is configured to generate the first reference voltage vreg_cdeg for thecapacitor controller DAC 220 using replica input FET 311, wherevreg_cdeg varies at least according to any variation in Vt of the inputFETs 241/242. In some embodiments, vreg_cdeg may further vary accordingto variations in common mode input signal Vcm. As the reference voltagevreg_cdeg varies according to PVT, it follows that an output voltage ofcapacitor controller DAC 220 responsive to a corresponding capacitorcontroller DAC code will vary accordingly, the output voltage providedto the gates of the varactors 221 and 222 of the CFET to maintaintemperature-tracked DAC linearity by keeping the CFET at a desiredcapacitive value despite e.g., temperature or common mode inputvariations.

Furthermore, the tail nodes 225/226 may be coupled together byconfiguring the RFET embodied by transistors 231 and 232. Whenconfigured by resistor controller DAC 230 to a low impedance, tail nodes225 and 226 are forced to the same voltage which is essentially afunction of voltage drop across current sources 210/211 and the voltageof 225 and 226 may vary with PVT according to e.g., temperature changescausing threshold voltage changes in the input FETs 241 and 242.Furthermore, when the RFET is configured by resistor controller DAC 230to a higher impedance, the RFET embodied by FETs 231 and 232 have agate-to-source voltage that is a function of both the DAC output voltageand the PVT-dependent tail node voltage. As shown in FIG. 3 thereference generation circuit generates a second reference voltagevreg_rdeg at the output of amplifier 360 that is provided to resistorcontroller DAC 230, where the second reference voltage vreg_rdeg varieswith respect to (i) PVT variations in the input FETs 241/242 (as thevoltage divider is operating according to the first reference voltagevreg_cdeg as a reference) and (ii) variations in threshold voltage ofFETs 231 and 232 that compose the RFET. Thus, the second referencevoltage vreg_rdeg similarly maintains temperature-tracked DAC linearityas any given DAC controller code provided to the resistor controller DACwill generate a corresponding RFET control voltage that variesaccordingly to maintain a desired resistive value of the RFET acrossPVT.

The design of capacitor controller DAC 220 and resistor controller DAC230 may follow conventional practices, with embodiments generallyincluding a reference voltage source, a resistive ladder used togenerate fractional parts or steps of said voltage, and selection of aparticular step as the output value according to a corresponding DACcode. Multiple forms of resistive ladders are known in the art,including R-2R, binary weighted, linear chain, etc. Similarly, resultselection may be controlled by binary, thermometer, or other controlsignal encoding, without limitation.

FIG. 3 is a schematic of one embodiment of a reference generationcircuit 300 generating first and second reference voltages vreg_cdeg andvreg_rdeg, respectively, compensating for PVT. In some embodiments, thereference generation circuit 300 may include a scaled replica 305 of oneleg of the actual CTLE circuit shown in FIG. 2 and may include currentsource 310 (which may correspond to a scaled replica of current sources210/211 in FIG. 2), replica input FET 311 (which may correspond to ascaled replica of input FETs 241/242), and cascode transistor 312 (whichmay be a scaled replica of cascode transistors 251/252). The actualvalues of CTLE load resistor RLO and inductor LO do not significantlyimpact the tail node voltage, so in the replica circuit they may bereplaced by the more compact load current sink 313, however nolimitation is implied. The quiescent voltages of the actual differentialcircuit are the same on both legs, so only one half of the circuit maybe duplicated in the reference generation circuit 300. The size andstructural parameters of the replica current source, input, and cascodetransistors may be identical to those of the actual CTLE instance ofFIG. 2 and, as with the actual CTLE circuit, the replica current sourcemay be biased by regulated voltage PBIAS, and the cascode transistor'sgate biased by regulated voltage PCASC. Thus, node voltages within thereference generation circuit track those of the actual CTLE overvariations in process, voltage, and temperature. In some lower-powerembodiments, the elements of the replica circuit may be a scaled versionof those in the CTLE circuit, in which a scaled current source may alsobe used to track the same voltage.

The CTLE circuit inputs include a differential signal superimposed on aDC bias voltage Vcm. In some embodiments the received input signals areAC coupled, and a local DC bias voltage Vcm is present at the CTLEinputs. The same local DC bias voltage, biases the replica input FET 311of the reference generation circuit 300. In embodiments in which thereceived input signals are DC coupled and thus the CTLE inputs arebiased by the actual input common-mode voltage, the input signals may besummed either passively with a resistor network or actively with a unitygain summing amplifier to provide Vcm to the reference generationcircuit 300.

Node 315 of the reference generation circuit is equivalent to that of atail circuit node 225/226 of the CTLE circuit of FIG. 2, thus the steadystate voltage of node 315 tracks that of the tail circuit nodes 225/226over PVT. Unity gain analog amplifier 320 buffers the voltage on node315 to produce the first reference voltage vreg_cdeg, which provides thereference level for e.g., the resistive ladder of capacitor controllerDAC 220 controlling the gates of varactors 221 and 222 of the CFET inthe receiver equalizer peaking circuit. As the capacitance of varactors221 and 222 are a function of their gate-to-channel voltage, (thechannel voltage being in this example the tail circuit voltage of nodes225 and 226), generating a reference voltage for capacitor controllerDAC 220 that varies according to PVT will provide voltage outputs to thegate control of varactors 221 and 222 that track the tail circuitvoltage, maintaining a near-constant voltage differential that minimizescapacitance variations for a given capacitor controller DAC code oradjustment value configuring capacitor controller DAC 220, regardless ofvariations of threshold voltage, supply voltage, or circuit operatingtemperature.

Reference voltage vreg_cdeg also powers the voltage divider 350 in thereference generation circuit 300 that tracks the variation in thethreshold voltage of FETs 231/232 in the RFET and the thermalcharacteristics of resistor controller DAC 230's resistive ladder,represented in the reference generation circuit by replica RFET 351 andpassive resistors 352, 353, 354, 355, 356. Specifically, replica RFET351 (and additionally but not necessarily required, resistor 352) offsetthe voltage at node 357 from vreg_cdeg by an amount proportional atleast to the threshold voltage Vt of the replica RFET 351. Similarly,for a given control code or adjustment value configuring resistor DAC220, the resulting resistor control voltage so derived from referencevoltage vreg_rdeg will track both (i) PVT variations of the voltages ontail nodes 225/226 (and therefore, variations in the threshold voltagesVt of input FETs 241 and 242) due to the voltage divider 350 operatingaccording to vreg_cdeg, as well as (ii) threshold voltage Vt variationsfor the RFET transistors 231, 232 via replica RFET 351, i.e., thereference voltage vreg_rdeg for the source degeneration switch composedof RFET transistors 231 and 232 tracks the threshold voltage of thesource degeneration switch itself.

As shown in FIG. 3, vreg_rdeg=vreg_cdeg−V_(gs,351) (the thresholdvoltage of replica RFET 351). As e.g., temperature changes, thethreshold voltage V_(t,351) of replica RFET 351 changes, and thus theV_(gs,351) also changes as V_(gs,351)=V_(ds,351)+V_(t,351).

To minimize PVT variations, replica RFET 351 may be designed to beidentical to FETs 231/232 of the RFET in the CTLE circuit of FIG. 2, andthe resistive elements 352-356 may be matched to those used in resistorcontroller DAC 230.

One tap on the resistor ladder 350 of reference regeneration circuit 300is buffered by unity gain analog amplifier 360 to produce referencevoltage vreg_rdeg, which provides the reference level for resistorcontroller DAC 230 controlling the resistance of transistors 231 and232. In some embodiments, resistor controller DAC 230 may include anequivalent structure as the resistor ladder 350 in the referencegeneration circuit. In one embodiment in accordance with FIGS. 2 and 3,resistor controller DAC 230 has an adjustment range between vreg_rdegand Vss, essentially spanning the gate voltages in which the RFET231/232 is in or near the active region, e.g. from cutoff to channelsaturation. In some embodiments as shown in FIG. 3, the voltage dividercircuit comprise at least one passive resistance element 352 between theoutput tap and the RFET 351, the passive resistive element 352 providinga voltage drop at least in part to increase a resolution of the resistorcontroller DAC. Utilizing resistor 352 in the reference generationcircuit 300 to match the DAC output range to the useful control rangefor transistor resistance may minimize the number of useless (e.g. outof range) DAC settings, thus allowing use of a resistor controller DAC230 having a higher resolution as the same number of codes may beavailable to resistor controller DAC 230, but with a lower total voltagerange. Alternative embodiments may simply implement fewer adjustmentsteps in a more compact implementation than a conventional full-rangeDAC producing an output that is utilized over only part of itsconfigurable range. In another embodiment, a different tap on theresistor ladder 350 of reference generation circuit 300 may be selectedto obtain a different range, or to accommodate a different DAC topology.

In another embodiment, the fixed current passing through replica inputFET 311 of the reference generation circuit is designed to be less thanthat of the actual CTLE circuit. In one particular embodiment, the proxyis operated at one fourth the differential circuit quiescent current, soas to reduce the overall standby current consumption of the system.

Just as [Rattan I] describes CTLE embodiments utilizing either PMOS orNMOS transistors, equivalent embodiments of the bias circuits describedherein may be based on NMOS transistors rather than the PMOS transistorsused in the present example, with the associated translation of supplyvoltages, reference voltages, and adjustment ranges understood to beassociated with such modification. Similarly, no limitation to a singletransistor type is implied, as further embodiments may incorporate mixedcombinations of PMOS and NMOS devices.

I claim:
 1. An apparatus comprising: a receiver equalizer peakingcircuit having a capacitor Field Effect Transistor (CFET) providing acapacitive value and a resistor FET (RFET) providing a resistive value;a controller circuit comprising a capacitor controller digital-to-analogconverter (DAC) configured to provide a CFET control voltage to the CFETbased on a first reference voltage, and a resistor controller DACconfigured to provide a RFET control voltage to a gate of the RFET basedon a second reference voltage; a reference generation circuit configuredto generate the first and second reference voltages, the referencegeneration circuit comprising: a replica input FET configured togenerate the first reference voltage that varies at least according to athreshold voltage (Vt) of an input FET, the replica input FET configuredto provide the first reference voltage to the capacitor controller DACto maintain temperature-tracked DAC linearity; a replica RFET configuredto generate the second reference voltage that varies with respect to (i)the first reference voltage and (ii) a Vt of the RFET, and to providethe second reference voltage to the resistor controller to maintain thetemperature-tracked DAC linearity.
 2. The apparatus of claim 1, whereinthe first reference voltage further varies responsive to variations in acommon mode voltage of an input signal received at the input FET.
 3. Theapparatus of claim 1, wherein the replica RFET is part of a voltagedivider circuit, and wherein the second reference voltage is generatedat an output tap of the voltage divider circuit.
 4. The apparatus ofclaim 3, wherein the replica RFET has a gate input connected to theoutput tap of the voltage divider circuit.
 5. The apparatus of claim 3,wherein the voltage divider circuit is a replica of a voltage divider inthe resistor controller DAC.
 6. The apparatus of claim 3, wherein thevoltage divider circuit comprise at least one passive resistance elementbetween the output tap and the RFET, the passive resistive elementproviding a voltage drop at least in part to increase a resolution ofthe resistor controller DAC.
 7. The apparatus of claim 1, wherein thereplica input FET is a scaled version of the input FET.
 8. The apparatusof claim 1, wherein the CFET control voltage and the RFET controlvoltage are variable output voltages that vary according to variationsin the first and second reference voltages, respectively.
 9. Theapparatus of claim 1, wherein the second reference voltage furthertracks the Vt of the input FET.
 10. The apparatus of claim 1, whereinthe CFET is connected in a varactor diode configuration, and wherein theCFET control voltage is provided to a gate of the CFET.
 11. A methodcomprising: equalizing an input signal according to a receiver equalizerpeaking circuit having a capacitor FET (CFET) providing a capacitivevalue and a resistor FET (RFET) providing a resistive value; generatinga capacitor control voltage at a gate of the CFET using a capacitorcontroller DAC based on a first reference voltage, and a RFET controlvoltage at a gate of the RFET using a resistor controller DAC based on asecond reference voltage; generating the first reference voltage using areplica input FET, the first reference voltage varying according to athreshold voltage (Vt) of an input FET; providing the first referencevoltage to the capacitor controller DAC to maintain temperature-trackedDAC linearity; generating the second reference voltage using a replicaRFET, the second reference voltage varying with respect to (i) the firstreference voltage and (ii) a Vt of the replica of the RFET; andproviding the second reference voltage to the resistor controller DAC tomaintain the temperature-tracked DAC linearity.
 12. The method of claim11, wherein the first reference voltage further varies responsive tovariations in a common mode voltage of an input signal received at theinput FET.
 13. The method of claim 11, wherein generating the secondreference voltage comprises generating a current through a voltagedivider circuit containing the replica RFET, wherein the current variesresponsive at least in part to variations in resistance of the replicaRFET, and wherein the second reference voltage is generated at an outputtap of the voltage divider circuit.
 14. The method of claim 13, whereinthe replica RFET has a gate input connected to the output tap of thevoltage divider circuit.
 15. The method of claim 13, wherein the voltagedivider circuit is a replica of a voltage divider in the resistorcontroller DAC.
 16. The method of claim 13, wherein the output tap ofthe voltage divider circuit is separated from the replica RFET by atleast one passive resistance element, the passive resistance elementgenerating a voltage drop at least in part for increasing a resolutionof the resistor controller DAC.
 17. The method of claim 11, wherein thereplica input FET is a scaled version of the input FET.
 18. The methodof claim 11, wherein the first reference voltage is process-dependentwith respect to the input FET.
 19. The method of claim 11, wherein thesecond reference voltage is process-dependent with respect to the inputFET and the RFET.
 20. The method of claim 11, wherein the CFET comprisesa varactor diode.